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1 ? fn9177.0 ISL6269 single phase pwm controller for mobile graphical processing unit (gpu) the ISL6269 ic is a single-phase synchronous-buck pwm controller featuring intersil's robust ripple regulator (r 3 ) technology that delivers truly superior dynamic response to input voltage and output load transients. integrated mosfet drivers, 5v ldo, and bootstrap diode result in fewer components and smaller implementation area. intersil?s r 3 technology combines the best features of fixed- frequency pwm and hysteretic pwm while eliminating many of their shortcomings. r 3 technology emplo ys an innovative modulator that synthesizes an ac ripple voltage signal v r , analogous to the output inductor ripple current. the ac signal v r enters a hysteretic comparator where the lower threshold is the error amplifier output v comp , and the upper threshold is a programmable voltage reference v w, resulting in generation of the pwm signal. the voltage reference v w sets the steady-state pwm frequency. both rising and falling edges of the pwm are modulated, providing faster response to input voltage transients and output load transients than conventional fixed-frequency pwm controllers. unlike a conventional hysteretic conver ter, the ISL6269 has an error amplifier that provides 1% voltage regulation at the fb pin. the ISL6269 has a 1.5ms digital soft-start and can be started into a pre-biased output voltage. a resistor divider is used to program the output voltage setpoint. the ISL6269 can be configured to operate in forced-continuous- conduction-mode (fccm) or in diode-emulation-mode (dem), which improves light-load efficiency. in fccm the controller always operates as a synchronous rectifier, switching the low-side mosfet regardless of the output load, however with dem enabled, the low-side mosfet is disabled preventing negative current flow from the output inductor during low load operation. an audio filter prevents the pwm switching frequency from entering the audible spectrum due to extremely light load while in dem. a pgood pin indicates when the converter is capable of supplying regulated voltage. t he ISL6269 features a unique fault-identification capability that can drastically reduce trouble-shooting time and effort. the pull-down resistance of the pgood pin is 30 ? for an overcurrent fault, 60 ? for an overvoltage fault, or 90 ? for either an undervoltage fault or during soft-start. the overcurrent protection is accomplished by measuring the voltage drop across the r ds(on) of the low-side mosfet. a single resistor programs the overcurrent and short-circuit points. overvoltage and undervoltage protection is monitored at the fb voltage feedback pin. features ? high performance synthetic ripple regulation ? extremely fast transient response ? external type-two loop compensation ? 1% regulation accuracy: -10c to +100c ? starts into a pre-biased output ? wide input voltage range: +7.0v to +25.0v ? wide output voltage range: +0.6v to +3.3v ? wide output load range: 0a to 25a ? programmable pwm frequency: 200khz to 600khz ? power good monitor ? fault identification by pgood pull down resistance ? integrated mosfet drivers with shoot-through protection ? internal digital soft-start ? internal 5v ldo regulator ? configure forced continuous conduction or diode emulation for increased light load efficiency ? pwm minimum frequency above audible spectrum ? integrated boot-strap diode ? low-side mosfet r ds(on) overcurrent protection ? undervoltage protection ? soft crowbar overvoltage protection ? over-temperature protection ? pb-free plus anneal available (rohs compliant) applications ? pci express graphical processing unit ? auxiliary power rail ?vrm ? network adapter data sheet june 14, 2005 caution: these devices are sensitive to electrosta tic discharge; follow proper ic handling procedures. 1-888-intersil or 1-888-468-3774 | intersil (and design) is a registered trademark of intersil americas inc. copyright intersil americas inc. 2005. all rights reserved all other trademarks mentioned are the property of their respective owners.
2 fn9177.0 june 14, 2005 pinout 16 ld qfn (4mm x 4mm) top view ordering information part number temp (c) package pkg. dwg. # ISL6269crz (see note) -10 to +100 16 ld 4x4 qfn (pb-free) l16.4x4 ISL6269crz-t (see note) 16 ld 4x4 qfn tape and reel (pb-free) l16.4x4 note: intersil pb-free plus anneal products employ special pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish , which are rohs compliant and compatible with both snpb and pb-free soldering operations. intersil pb-free products are msl classi fied at pb-free peak reflow temperatures that meet or exceed the pb-free requirements of ipc/jedec j std-020. 1 3 4 15 16 14 13 2 12 10 9 11 6 578 gnd boot pgood ug phase vo comp fset fb isen pvcc pgnd lg en vin fccm vcc ISL6269
3 fn9177.0 june 14, 2005 absolute voltage ratings isen, vin to gnd. . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3v to +28v vcc, pgood to gnd . . . . . . . . . . . . . . . . . . . . . . . . -0.3v to +7.0v pvcc to pgnd. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3v to +7.0v gnd to pgnd. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3v to +0.3v en, fccm. . . . . . . . . . . . . . . . . . . . . . . . -0.3v to gnd, vcc +3.3v phase to gnd. . . . . . . . . . . . . . . . . . . . . . . . . . (dc) -0.3v to +28v . . . . . . . . . . . . . . . . . . . . . . . . . . (<100ns pulse width, 10 j) -5.0v boot to gnd, or pgnd . . . . . . . . . . . . . . . . . . . . . . -0.3v to +33v boot to phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3v to +7v ug. . . . . . . . . . . . . . . . . . . . . . . (dc) -0.3v to phase, boot +0.3v . . . . . . . . . . . . . . . . . . . . . . . . . . (<200ns pulse width, 20 j) -4.0v lg . . . . . . . . . . . . . . . . . . . . . . . . (dc) -0.3v to pgnd, pvcc +0.3v . . . . . . . . . . . . . . . . . . . . . . . . . . . (<100ns pulse width, 4 j) -2.0v esd classification . . . . . . . . . . . . . . . . . . . . . .level 1 (hbm = 2kv) thermal information thermal resistance (typical, notes 1, 2) ja (c/w) jc (c/w) qfn package . . . . . . . . . . . . . . . . . . . 43 11.5 junction temperature range . . . . . . . . . . . . . . . . . -55 c to +150 c operating temperature range . . . . . . . . . . . . . . . . -10 c to +100 c storage temperature . . . . . . . . . . . . . . . . . . . . . . . -65 c to +150 c lead temperature . . . . . . . . . . . . . . . . . . . . (soldering, 10s)+300 c recommended operating conditions ambient temperature range . . . . . . . . . . . . . . . . . . -10c to 100c supply voltage (vin to gnd) . . . . . . . . . . . . . . . . . . . . . . 7v to 25v caution: stress above those listed in ?absolute maximum ratings ? may cause permanent damage to the device. this is a stress onl y rating and operation of the device at these or any other conditions above those indicated in th e operational section of this specification is not implied. note: 1. ja is measured with the component mounted on a highly effective t hermal conductivity test board on free air. see tech brief tb379 for details 2. for jc , the ?case temp? location is the center of the exposed metal pad on the package underside. electrical specifications these specifications apply for v in = 15v, t a = (-10c) to (+100c), unless otherwise stated. all typical spec ifications t a = (+25c) parameter symbol test conditions min typ max unit vin vin voltage range v in 7.0 25.0 v vin input bias current i vin en and fccm = 5v, fb = 0.65v, vin = 7v to 25v 2.2 3.0 ma vin shutdown current i shdn en = gnd, vin = 25v 0.1 1.0 a vcc ldo vcc output voltage range v cc vin = 7v to 25v, i ldo = 0ma to 80ma 4.75 5.00 5.25 v vcc por threshold rising vcc por threshold voltage v ccthr 4.35 4.45 4.55 v falling vcc por threshold voltage v ccthf 4.10 4.20 4.30 v regulation error amplifier reference voltage v ref 0.6 v voltage regulation accuracy v reg -1 +1 % pwm frequency range f osc fccm = 5v 200 600 khz f audio fccm = gnd 21 28 khz frequency-set accuracy f osc = 300khz -12 +12 % vo range v vo 0.60 3.30 v vo input leakage current i vo vo = 0.60v vo = 3.30v 1.3 7.0 a a error amplifier fb input bias current i fb fb = 0.60v 20 na comp source current i compsrc fb = 0.40v, comp = 3.20v 2.5 ma comp sink current i compsnk fb = 0.80v, comp = 0.30v 0.3 ma comp high clamp voltage v comphc fb = 0.40v, sink 50 a 3.10 3.40 3.65 v comp low clamp voltage v complc fb = 0.80v, source 50 a 0.09 0.15 0.21 v ISL6269
4 fn9177.0 june 14, 2005 power good pgood pull down impedance pgr ss pgood = 5ma sink 80 95 133 ? pgr uv pgr ov pgood = 5ma sink 53 63 89 ? pgr oc pgood = 5ma sink 26 32 46 ? pgood leakage current i pgood pgood = 5v <0.1 1.0 a pgood maximum sink current 5.0 ma pgood soft-start delay t ss en high to pgood high 2.20 2.75 3.30 ms gate driver ug pull-up resistance r ugpu 200ma source current (note 2) 1.0 1.5 ? ug source current i ugsrc v ug to phase = 2.5v 2.0 a ug sink resistance r ugpd 250ma sink current (note 2) 1.0 1.5 ? ug sink current i ugsnk v ug to phase = 2.5v 2.0 a lg pull-up resistance r lgpu 250ma source current (note 2) 1.0 1.5 ? lg source current i lgsrc v lg to pgnd = 2.5v 2.0 a lg sink resistance r lgpd 250ma sink current (note 2) 0.5 0.9 ? lg sink current i lgsnk v lg to pgnd = 2.5v 4.0 a delay from ug falling to lg rising t ugflgr ug falling to lg rising 21 ns delay from lg falling to ug rising t lgfugr lg falling to ug rising 14 ns bootstrap diode forward voltage v f pvcc = 5v, i f = 2ma 0.58 v reverse leakage i r v r = 25v 0.2 a control inputs en high threshold voltage v enthr 2.0 v en low threshold voltage v enthf 0.5 v fccm high threshold voltage v fccmthr 2.0 v fccm low threshold voltage v fccmthf 1.0 v en leakage current i enl en = 0v <0.1 1.0 a i enh en = 5.0v 20 a fccm leakage current i fccml fccm = 0v <0.1 1.0 a i fccmh fccm = 5.0v 2.0 a protection isen ocp threshold current i oc -33 -26 -19 a isen short-circuit threshold current i sc -50 a uvp threshold voltage v uv 81 84 87 % ovp rising threshold voltage v ovr 113 116 119 % ovp falling threshold voltage v ovf 103 % otp rising threshold temperature t otr (note 2) 150 c otp temperature hysteresis t othys (note 2) 25 c note: 3. guaranteed by design. electrical specifications these specifications apply for v in = 15v, t a = (-10c) to (+100c), unless otherwise stated. all typical spec ifications t a = (+25c) (continued) parameter symbol test conditions min typ max unit ISL6269
5 fn9177.0 june 14, 2005 functional pin descriptions gnd pin bottom terminal pad of qfn package signal common of the ic. unless otherwise stated, signals are referenced to the g nd pin, not the pgnd pin. vin pin-1 (input) the vin pin measures the converter input voltage with respect to the gnd pin. vin is a required input to the r 3 pwm modulator. the vin pin is al so the input source for the integrated +5v ldo regulator. vcc pin-2 (output) the vcc pin is the output of the integrated +5v ldo regulator, which provides the bias voltage for the ic. the vcc pin delivers regulated +5v whenever the en pin is pulled above v enthr . for best performance the ldo requires at least a 1f mlcc decouple capacitor to the gnd pin. fccm pin-3 (logic) the fccm pin configures the cont roller to operate in forced- continuous-conduction-mode (fccm) or diode-emulation- mode (dem.) dem is disabled when the fccm pin is pulled above the rising threshold voltage v fccmthr , and dem is enabled when the fccm pin is pulled below the falling threshold voltage v fccmthf. en pin-4 (logic) the en pin is the on/off switch of the ic. when the en pin is pulled above the rising threshold voltage v enthr, v cc will ramp up and begin regulation. the soft-start sequence begins once v cc ramps above the power-on reset (por) rising threshold voltage v ccthr . when the en pin is pulled below the falling threshold voltage v enthf , pwm immediately stops and v cc decays below the por falling threshold voltage v ccthf , at which time the ic turns off. comp pin-5 (signal) the comp pin is the output of the control-loop error amplifier. loop compensation components connect from the comp pin to the fb pin. fb pin-6 (signal) the fb pin is the inverting input of the control loop error amplifier. the converter will regulate to 600mv at the fb pin with respect to the gnd pin. scale the desired output voltage to 600mv with a voltage divider network made from resistors r top and r bottom . loop compensation components connect from the fb pin to the comp pin. fset pin-7 (signal) the fset pin programs the pwm switching frequency of the converter. connect a resistor r fset and a 10nf capacitor c fset from the fset pin to the gnd pin. vo pin-8 (input) the vo pin makes a direct measurement of the converter output voltage used exclusively by the r 3 pwm modulator. the vo pin should be connected to the top of feedback resistor r top at the converter output. refer to figure 1, typical application schematic. isen pin-9 (input) the isen pin is the input to th e overcurrent protection (ocp) and short-circuit protection (scp) circuits. connect a resistor r sen between the isen pin and the phase pin. select the value of r sen that will force the isen pin to source the i sen threshold current i oc when the peak inductor current reaches the desired ocp setpoint. the scp threshold current i sc is fixed at twice the ocp threshold current i oc pgnd pin-10 the pgnd pin should be connected to the source of the low- side mosfet, preferably with an isolated path that is in parallel with the trace connecting the lg pin to the gate of the mosfet. the pgnd pin is an isolated path used exclusively to conduct the turn-off transient current that flows out the pgnd pin, through the gate-source capacitance of the low-side mosfet, into the lg pin, and back to the pgnd pin through the pull-down resistance of the lg driver. the adaptive shoot-through prot ection circuit, measures the low-side mosfet gate voltage with respect to the pgnd pin, not the gnd pin. lg pin-11 (output) the lg pin is the output of the low-side mosfet gate driver. connect to the gate of the low-side mosfet. pvcc pin-12 (input) the pvcc pin is the input voltage for the low-side mosfet gate driver lg. connect a +5v power source to the pvcc pin with respect to the gnd pin, a 1f mlcc bypass capacitor needs to be connected from the pvcc pin to the pgnd pin, not the gnd pin. the vcc output may be used for the pvcc input voltage source. connect the vcc pin to the pvcc pin through a low-pass filter consisting of a resistor and the pvcc bypass capacitor. refer to figure 1, typical application schematic . ISL6269
6 fn9177.0 june 14, 2005 boot pin-13 (input) the boot pin is the input voltage for the high-side mosfet gate driver ug. an mlcc capacitor c boot is connected between the boot pin and the phase pin, the return current path for the ug mosfet driver. capacitor c boot is charged from the voltage sour ce at the pvcc pin via the internal diode d boot each time the phase pin drops below pvcc minus diode d boot forward voltage drop v f . ug pin-14 (output) the ug pin is the output of the high-side mosfet gate driver. connect to the gate of the high-side mosfet. phase pin-15 (input) the phase pin is the return current path for the ug mosfet driver. the phase pin also measures the polarity of the low-side mosfet drain voltage for the diode emulation function. connect the phase pin to the node consisting of the high-side mosfet source, the low-side mosfet drain, and the output inductor. refer to figure 1, typical application schematic . pgood pin-16 (output) the pgood pin is an open-drain output that is high impedance when the converter is in regulation, or when the en pin is pulled below the falling threshold voltage v enthf . the pgood pin has three distinct pull-down impedances that correspond to an ovp fault, ocp/scp, or uvp and soft- start. connect the pgood pin to a pull-up resistor. ISL6269
7 fn9177.0 june 14, 2005 typical application figure 1. typical application schematic vcc pvcc fset gnd en fccm pgood comp fb vin phase isen boot ug lg pgnd c pvcc c out l out ISL6269 vo c vcc r pvcc r pgood q high_side r sen r fset c fset r top r bottom r comp c comp1 c comp2 c boot vin c in v out 0.6v-3.3v q low_side 7v-25v ISL6269
8 fn9177.0 june 14, 2005 block diagram figure 2. schematic block diagram v w v r pwm r pgood en fb comp isen boot ug lg pvcc pgnd fccm phase uvp s q por digital soft-start 150ot shoot through protection driver driver v comp pwm control vin fset vcc vo v ref ldo + ? ovp + ? ea ? + ea ocp + ? ? + g m v in ? + g m v o ? + ? + ? + c r pwm frequency control gnd package bottom i oc 30 ? 90 ? 60 ? ? + ISL6269
9 fn9177.0 june 14, 2005 theory of operation modulator the ISL6269 is a hybrid of fixed frequency pwm control, and variable frequency hysteretic control. the term ?ripple? in the name ?robust-ripple-regulator? refers to the converter output inductor ripple current, not the converter output ripple voltage. the output voltage is regulated to 600mv at the fb pin with respect to the gnd pin. the fb pin is the inverting input of the error amplifier. the frequency response of the feedback control loop is tuned with a type-two compensation network connected across the fb pin and comp pin. the r 3 modulator synthesizes an ac signal v r , which is an ideal representation of the output inductor ripple current. the duty-cycle of v r is derived from the voltage measured at the vin pin and vo pin with respect to the gnd pin. transconductance amplifiers convert the vin and vo voltages into currents that charge and discharge the ripple capacitor c r . the positive slope of v r can be written as: the negative slope of v r can be written as: a voltage v w is referenced with respect to the error amplifier output voltage v comp , creating a window-voltage envelope into which voltage v r is compared. the v r, v comp, and v w signals feed into a hysteretic window comparator in which v comp is the lower threshold voltage and v w is the higher threshold voltage. pwm pulses are generated as v r traverses the v w and v comp thresholds. the charging and discharging rates of capacitor c r determine the pwm switching frequency for a given amplitude of v w with respect to v comp . the r 3 regulator simultaneously affects switching frequency and duty cycle because it modulates both edges of the pwm pulses. v rpos gm () v in v o ? () ? = (eq. 1) v rneg gm v o ? = (eq. 2) figure 3. modulator waveforms during load transient s volts ripple capacitor voltage c r pwm error amplifier voltage v comp window voltage v w ISL6269
10 fn9177.0 june 14, 2005 ldo voltage applied to the vin pin with respect to the gnd pin is regulated to +5vdc by an in ternal low-dropout voltage regulator (ldo). the output of the ldo is called v cc , which is the bias voltage used by the ic internal circuitry. the ldo output is routed to the vcc pin and requires a ceramic capacitor connected to the gnd pin to stabilize the ldo and to decouple load transients. when the en pin rises above the v enr threshold, v cc will turn on and rise to its regulation voltage. the ldo regulates v cc by pulling up towards the voltage at the vin pin; the ldo has no pull-down capability. por and soft-start the power-on reset (por) circuit monitors v cc for the v ccr (rising) and v ccf (falling) voltage thresholds. the purpose of soft-start is to lim it the inrush current through the output capacitors when the converter first turns on.the pwm soft-start sequence initializes once v cc rises above the v ccr threshold, beginning from below the v ccf threshold. the ISL6269 uses a digital soft-start circuit to ramp the output voltage of the converter to the programmed regulation setpoint in approximately 1.5m s. the converter regulates to 600mv at the fb pin with respect to the gnd pin. during soft-start a digitally derived voltage reference forces the converter to regulate from 0v to 600mv at the fb pin. when the en pin is pulled below the v enf threshold, the ldo stops regulating and pwm immediately stops, regardless of the falling v cc voltage. the soft-start sequence can be reinitialized and fault latches reset, once v cc falls below the v ccf threshold. mosfet gate-drive outputs the ISL6269 incorporates a mosfet driver that controls both high-side and low-side n-channel mosfets. the drivers are optimized for lo w duty-cycle applications prevalent with large step down voltages. at low duty-cycle, the low-side mosfet conducts for a much longer time in a switching period than the high-side mosfet, necessitating lower r ds(on) at the expense of larger parasitic capacitance. the low-side gate driver is th erefore sized much larger to meet this application requirem ent. the larger sink current capability enables the low-side gate driver to hold the gate- source voltage of the mosfet below its v gsth as current conducts through the drain-to-gate parasitic capacitance. both drivers incorporate ada ptive shoot-through protection to prevent high-side and low-side mosfets from conducting simultaneously and shorting the input supply. during turn-off of the low-side mosfet, the lg to pgnd voltage is monitored until it reaches a 1v threshold, at which time the ug driver is allowed to switch. during turn-off of the high-side mosfet, the ug to phase voltage is monitored until it reaches a 1v threshold, at which time the lg driver is allowed to switch. the input power for the lg driver circuit is sourced directly from the pvcc pin. the input power for the ug driver circuit is sourced from a ?boot? capacitor connected from the boot pin to the phase pin. the same supply that is connected to the pvcc pin is used to charge the boot capacitor via the internal schottky diode of the ic. ug lg t lgfugr t ugflgr figure 4. gate drive timing diagram ISL6269
11 fn9177.0 june 14, 2005 diode emulation positive inductor current can flow from the source of the high-side mosfet or from the drain of the low-side mosfet. negative inductor current flows into the drain of the low-side mosfet. when the low-side mosfet conducts positive inductor current, the phase voltage will be negative with respect to the gn d pin. conversely, when the low-side mosfet conducts negative inductor current, the phase voltage will be positive with respect to the gnd pin. negative inductor current occurs when the output load current is less than ? the inductor ripple current. the ISL6269 can be configured to operate in forced- continuous-conduction-mode (fccm) or in diode-emulation- mode (dem), which can improve light-load efficiency. in fccm, the controller always operates as a synchronous rectifier, switching the low-side mosfet regardless of the polarity of the output inductor current. in dem, the low-side mosfet is disabled during negat ive current flow from the output inductor. dem is permitted when the fccm pin is pulled low, and disabled when pulled high. when dem is permitted, the converter will automatically select fccm or dem according to load conditions. if positive phase pin voltage is measured for eight co nsecutive pwm pulses, then the converter will enter diode-emulation mode on the next pwm cycle. if a negative phase pin voltage is measured, the converter will ex it dem on the following pwm pulse. an audio filter is incorporated into the pwm generation circuitry that pr events the switching frequency from entering the audible spectrum at low load conditions. overcurrent and short-circuit protection when an ocp or scp fault is detected, the ISL6269 overcurrent and short-circuit protection circuit will pull the pgood pin low and latch off the converter. the fault will remain latched until the en pin is pulled below v enf or if the voltage at the vin pin is reduced to the extent that v cc has fallen below the por v ccf threshold. selecting the appropriate value of resistor r sen programs the ocp threshold. the resistor r sen is connected from the isen pin to the phase pin. the phase pin is connected to the drain terminal of the low-side mosfet. the ocp circuit measures positive-flowing, peak-current through the output inductor, no t the dc current flowing from the converter to the load. the low-side mosfet drain current is assumed to be equa l to the positive output inductor current when the high-side mosfet is turn off. current briefly conducts through the low-side mosfet body diode until the lg driver goe s high. the peak inductor current develops a voltage across the r ds(on) of the low- side mosfet just as if it were a discrete current-sense resistor. an ocp fault will occur when the isen pin has measured more than the ocp threshold current i oc, on consecutive pwm pulses, for a period exceeding 20s. it does not matter how many pwm pulses are measured during the 20s period. if a measurement falls below i oc before 20s has elapsed, then the timer is reset to zero. an scp fault will occur when the isen pin has measured more than the short-circuit threshold current i sc, in less than 10s, on consecutive pwm pulses. the relationship between i d and i sen can be written as: the value of rsen can then be written as: where: -r sen ( ? ) is the resistor used to program the over- current setpoint -i sen is the current sense current that is sourced from the isen pin -i oc is the i sen threshold current value sourced from the isen pin that will activate the ocp circuit -i fl is the maximum continuous dc load current -i pp is the inductor peak-to-peak ripple current -oc sp is the desired overcurrent setpoint expressed as a multiplier relative to i fl overvoltage when an ovp fault is detect ed, the ISL6269 overvoltage protection circuit will pull the pgood pin low and latch off the converter. the fault will remain latched until the en pin is pulled below v enf or if the voltage at the vin pin is reduced to the extent that v cc has fallen below the por v ccf threshold. when the voltage at the fb pin relative to the gnd pin, has exceeded the rising overvoltage threshold v ovr , the converter will latch off however, the lg driver output will stay high, forcing the low-side mosfet to pull down the output voltage of the converter. the low-side mosfet will continue to pull down the output voltage until the voltage at the fb pin relative to the gnd pin, has decayed below the falling overvoltage threshold v ovf, at which time the lg driver output is driven low, forcing the low-side mosfet off. the lg driver output will cont inue to switch on at v ovr and switch off at v ovf until the en pin is pulled below v enf or if the voltage at the vin is r educed to the extent that v cc has fallen below the por v ccf threshold. i ? sen r sen ? i ? d r ds on () ? = (eq. 3) r sen i fl i pp 2 -------- - + oc sp ? r ds on () ? i oc ---------------------------------------------------------------------------- = (eq. 4) ISL6269
12 fn9177.0 june 14, 2005 undervoltage when an uvp fault is detected, the ISL6269 undervoltage protection circuit will pull t he pgood pin low and latch off the converter. the uvp fault occurs when the voltage at the fb pin relative to the gnd pin, has fallen below the under- voltage threshold v uv. the fault will remain latched until the en pin is pulled below v enf or if the voltage at the vin is reduced to the extent that v cc has fallen below the por v ccf threshold. over-temperature when an otp fault is detected, the ISL6269 over- temperature protection circui t suspends pwm, but will not affect the pgood pin, or latch off the converter. the over- temperature protection circuit measures the temperature of the silicon and activates when the rising threshold temperature t otr has been exceeded. the pwm remains suspended until the silicon temperature falls below the temperature hysteresis t othys at which time normal operation is resumed. all other protection circuits will function normally during otp however, since pwm is inhibited, it is likely that the converter will immediately experience an undervoltage fault, latch off, and pull pgood low. if the en pin is pulled below v enf or if the voltage at the vin is reduced to the extent that v cc has fallen below the por v ccf threshold, normal operation will resume however, the temperat ure hysteresis t othys is reset. pgood the pgood pin connects to three open drain mosfets each of which has a different r ds(on) . consult the electrical specifications table for the pull-down resistance of pgood for the corresponding fault. the pgood pin is high impedance whenever v cc is below the rising por threshold v ovr , the falling por threshold v ovf , after delay t ss elapses, without an ovp, ocp, scp, or uvp fault. this fault- identification capability is a useful tool for trouble-shooting. component selection programming the output voltage when the converter is in regulation there will be 600mv from the fb pin to the gnd pin. connect a two-resistor voltage divider across the vo pin and the gnd pin with the output node connected to the fb pin. scale the voltage-divider network such that the fb pi n is 600mv with respect to the gnd pin when the converter is regulating at the desired output voltage. programming the output vo ltage can be written as: where: -v out is the desired output voltage of the converter. -v ref is the voltage that the converter regulates to at the fb pin. -r top is the voltage-programming resistor that connects from the fb pin to the vo pin. it is usually chosen to set the gain of the control-loop error amplifier. it follows that r bottom will be calculated based upon the already selected value of r top. -r bottom is the voltage-programming resistor that connects from the fb pin to the gnd pin. calculating the value of r bottom can now be written as: programming the pwm switching frequency the pwm switching frequency f osc is programmed by the resistor r fset that is connected from the fset pin to the gnd pin. programming the approximate pwm switching frequency can be written as: estimating the value of r fset can now be written as: where: -f osc is the pwm switching frequency. -r fset is the f osc programming resistor. - 60 x [1 x 10 -12 ] is a constant. selection of the lc output filter the duty cycle of a buck converte r is ideally a f unction of the input voltage and the output volt age. this relationship can be written as: where: - d is the pwm duty cycle. -v in is the input voltage to be converted. -v out is the regulated output vo ltage of the converter. the output inductor peak-to-peak ripple current can be written as: table 1. pgood pull-down resistance condition pgood resistance ic off open soft start 95 ? undervoltage fault 95 ? overvoltage fault 60 ? overcurrent fault 30 ? v ref v out r bottom r top r bottom + --------------------------------------------------- ? = (eq. 5) r bottom v ref r ? top v out v ref ? ------------------------------------- = (eq. 6) f osc 1 60 r ? fset 1 12 ? 10 [] ? ----------------------------------------------------------- = (eq. 7) r fset 1 60 f osc 1 12 ? 10 [] ? ? -------------------------------------------------------- = (eq. 8) dv in () v out v in --------------- - = (eq. 9) i pp v out 1dv in () ? [] ? f osc l o ? ---------------------------------------------------- - = (eq. 10) ISL6269
13 fn9177.0 june 14, 2005 where: -i pp is the peak-to-peak output inductor ripple current. -f osc is the pwm switching frequency. -l o is the nominal value of the output inductor. a typical step-down dc/dc converter will have an i pp of 20% to 40% of the nominal dc output load current. the value of i pp is selected based upon se veral criteria such as mosfet switching loss, indu ctor core loss, and the resistance the inductor winding , dcr. the dc copper loss of the inductor can be estimated by: the inductor copper loss can be significant in the total system power loss. attention has to be given to the dcr selection. another factor to consider when choosing the inductor is its saturation characteristics at elevated temperature. a saturated induct or could cause destruction of circuit components, as well as nuisance ocp faults. a dc/dc buck regulator must have output capacitance c o into which ripple current i pp can flow. current i pp develops a corresponding ripple voltage v pp across c o, which is the sum of the voltage drop across the capacitor esr and of the voltage change stemming from charge moved in and out of the capacitor. these two voltages can be written as: and if the output of the converter has to support a load with high pulsating current, several capacitors will need to be paralleled to adjust the esr to achieve the required v pp . the inductance of the capacitor can cause a brief voltage dip when the load transient has an extremely high slew rate. low inductance capacitors constructed with reverse package geometry are available. a capacitor dissipates heat as a function of rms current. be sure that i pp is shared by a sufficient quantity of paralleled capacitors so that they operate below the maximum rated rms current. take into account that the specified value of a capacitor can drop as much as 50% as the dc voltage across it increases. selection of the input capacitor the important parameters for the bulk input capacitance are the voltage rating and the rms current rating. for reliable operation, select bulk capacitors with voltage and current ratings above the maximum input voltage and capable of supplying the rms current requir ed by the switching circuit. their voltage rating should be at least 1.25 times greater than the maximum input voltage, while a voltage rating of 1.5 times is a preferred rating. fo r most cases, the rms current rating requirement for the input capacitors of a buck regulator is approximately 1/2 the dc output load current. the maximum rms current required by the regulator can be approximated through the following equation: where: - d is the converter duty cycle. -i rms is the input capacitance rms ripple current. -i load is the converter output dc load current. in addition to the bulk capacitance, some low esl ceramic capacitance is recommended to decouple between the drain terminal of the high-side mosfet and the source terminal of the low-side mosfet, in order to reduce the voltage ringing created by the switch ing current across parasitic circuit elements. mosfet selection and considerations typically, mosfets cannot tolerate even brief excursions beyond their maximum drain to source voltage rating. the mosfets used in the power conversion stage of the converter should have a maximum v ds rating that exceeds the upper voltage tolerance of the input power source, and the voltage spike that occurs when the mosfet switches off. placing a low esr ceramic capacitor as close as practical across the drain of the high-side mosfet and the source of the low-side mosfet will reduce the amplitude of the turn-off voltage spike. the mosfet input capacitance c iss, and on-state drain to source resistance r ds(on) , are to an extent, inversely related; reduction of r ds(on) typically results in an increase of c iss . these two parameters affect the efficiency of the converter in different ways. the r ds(on) affects the power loss when the mosfet is completely turned on and conducting current. the c iss affects the power loss when the mosfet is actively switching. switching time increases as c iss increases. when the mosfet switches it will briefly conduct current while the drain to source voltage is still present. the power dissipation during this time is substantial so it must be kept as short as practical. often the high-side mosfet and the low-side mosfet are different devices due to the trade-offs that have to be made between c iss and r ds(on) . the low-side mosfet power loss is dominated by r ds(on) because it conducts current for the majority of the pwm switching cycle; the r ds(on) should be small. the switching loss is small for the low-side mosfet even though c iss is large due to the low r ds(on) of the device, because the drain to source voltage is clamped by the body diode. the high- side mosfet power loss is dominated by c iss because it conducts current for the minority of the pwm switching cycle; the c iss should be small. the switching loss of the high-side mosfet is large compared to the low-side mosfet because the drain to source voltage is not clamped. for the lower mosfet, its power loss can be p copper i load [] 2 dcr ? = (eq. 11) ? v esr i pp e ? sr = (eq. 12) ? v c i pp 8c o f ? osc ? ---------------------------------- - = (eq. 13) i rms i load d [] d [] 2 ? ? = (eq. 14) ISL6269
14 fn9177.0 june 14, 2005 assumed to be the conduction loss only and can be written as: for the high-side mosfet, its conduction loss can be written as: for the high-side mosfet, its switching loss can be written as: the peak and valley current of the inductor can be obtained based on the inductor peak-to-peak current and the load current. the turn-on and turn-off time can be estimated with the given gate driver parameters in the electrical specification table. selecting the bootstrap capacitor the selection of the bootstrap capacitor can be written as: where: -q g is the total gate charge required to switch the high- side mosfet - ? v boot , is the maximum allowed voltage decay across the boot capacitor each time the mosfet is switched on as an example, suppose the high-side mosfet has a total gate charge q g , of 25nc at v gs = 5v, and a ? v boot of 200mv. the calculated bootstrap capacitance is 0.125f; select at least the first standa rd component value of greater capacitance than calculated, that being 0.15f. use an x7r or x5r ceramic capacitor. layout considerations power and signal layer placement on the pcb as a general rule, power layers should be adjacent to one another towards one side of the board, with signal layers adjacent to one another towards the opposite side of the board. for example, prospective layer arrangement on a 4 layer board is shown below: 1. top layer: ISL6269 signal lines 2. signal ground 3. power layers: power ground 4. bottom layer: power mosfet, inductors and other power traces it is a good engineering practice to separate the power conductors from the signal co nductors. the controller ic will stay on the signal layer, which is isolated by the signal ground to the power signal traces. the loop formed by the bottom mosfet, output indu ctor, and output capacitor, should be very small. a guard-ring placed around high impedance inputs fb and fset is recommended. component placement power mosfets should be placed close to the ic so that vin, lg, ug, phase, boot, a nd isen traces can be short. place components in such a way that the area near the fset, fb, comp, and vo pins avoid traces with high dv/dt and di/dt, such as gate signals and phase node signals. signal ground and power ground connection the bottom of the ISL6269 qfn package is the analog and logic ground terminal (gnd) of the ic. connect the gnd pad of the ISL6269 to the signal ground layer of the pcb using at least five vias, for a robust thermal and electrical conduction path. the best tie-point between the signal ground and the power ground is at the negativ e side of the output capacitors that is not in the return path of the inductor ripple current flowing through the output capacitors. pin 1 (vin) the vin pin should be connec ted to the drain of the high- side mosfet, using a low resistance and low inductance path. pin 2 vcc for best performance the ldo requires at least a 1f mlcc decouple capacitor connected from the vcc pin to the gnd pin. pin 3 (fccm) and pin 4 (en) these are logic inputs that are referenced to the gnd pin. treat as a typical logic signal. p conls dv in () ? i load [] 2 r ? ds on () ls 1dv in () ? [] ? (eq. 15) p conhs dv in () ? i load [] 2 r ? ds on () hs d ? v in () = (eq. 16) p swhs v in () v in i val t on f ? osc ? ? 2 ------------------------------------------------------------- v in i peak t off f ? osc ? ? 2 -------------------------------------------------------------------- - + = (eq. 17) c boot q g ? v boot ----------------------- - = (eq. 18) figure 5. typical power component placement v in - + v - + l o v in - + o - + l o ISL6269
15 fn9177.0 june 14, 2005 pin 5 (comp) the loop compensation components connect from the comp pin to the fb pin. place the components close to the fb pin to make the traces as short as possible. pin 6 (fb) there is usually a resistor divider connecting the output voltage of the converter to the fb pin. the correct layout should bring the output voltage from the regulation point to the fb pin with kelvin traces. the input impedance of the fb pin is high, so place the resistor divider close to the pin, keeping the high impedance trace short. pin 7 (fset) this pin requires a quiet environment. the resistor r fset and capacitor c fset should be placed directly adjacent to this pin. keep fast moving nodes away from this pin. pin 8 (vo) the vo pin should be connected to the kelvin traces at the fb voltage divider. pin 9 (isen) the isen trace should be routed away from the traces and components connected to the fb pin, comp pin, and fset pin. pin 10 (pgnd) this is the pull-down return path for the lg low-side mosfet gate drive. this should be an isolated low- resistance, low-inductance trace that connects to the source of the low-side mosfet. pin 11 (lg) connect to the gate terminal of the low-side mosfet. the signal going through this trace is both high dv/dt and high di/dt, with high peak charging and discharging current. route this trace in parallel with the trace from the pgnd pin. these two traces should be short, wide, and away from other traces. there should be no other weak signal traces in parallel with these traces on any layer. pin 12 (pvcc) a ceramic decoupling capacitor connects from the pvcc pin to the pgnd pin, not the gnd pin. closely place the capacitor on the same side of the board as the ISL6269 ic. pin 13 (boot) the di/dt and dv/dt of this pin are as high as that of the lg pin, ug pin, and the phase pin; therefore, t he traces should be as short as possible. pin 14 (ug) connect to the gate terminal of the high-side mosfet. the signal going through this trace is both high dv/dt and high di/dt, with high peak charging and discharging current. route this trace in para llel with the trace from the phase pin. these two traces should be short, wide, and away from other traces. there should be no other weak signal traces in parallel with these traces on any layer. pin 15 (phase) connect to the low-side mosfet drain terminal. the phase node has a very high dv/dt with a voltage swing from the input voltage to ground. this trace should be short, and positioned away from other weak signal traces. pin 16 (pgood) a very robust pin. treat as a typical logic signal. copper size for the phase node the parasitic capacitance and parasitic inductance of the phase node should be kept very low to minimize ringing. if ringing is excessive, it could easily affect current sample information. it would be best to limit the size of the phase node copper in strict accordance with the current and thermal management of the application. identify the power and signal ground the input and output capacitors of the converter, the source terminal of the low-side mosfet, and the pgnd pin should be closely connected to the power ground. the other components should connect to signal ground. signal and power ground are tied together at the negative terminal of the output capacitors. decoupling capacitor for switching mosfet ceramic capacitors should be closely connected to the drain side of the high-side mosfet, and the source of the low- side mosfet. this capacitor reduces the amplitude of the turn-off voltage spike. control loop the control loop model of th e ISL6269 is partitioned into function blocks consisting of: - the duty cycle to vo transfer function g vd (s) which is determined by the value of the output power components, input voltage, and output voltage. - the vcomp to duty cycle transfer function f m (s) which is determined by the pwm frequency, input voltage, output voltage, resistor r fset , and capacitor c fset. - the product of the g vd (s) and f m (s) transfer functions is expressed as the vcomp to vo transfer function g vovc (s). - the type-two compensation network g comp (s) that connects across the comp and fb pins. - the product of the g comp (s) and g vovc (s) transfer functions is expressed as the loop transfer function t(s). ISL6269
16 fn9177.0 june 14, 2005 the compensator zero fz1 is written as: the compensator pole fp1 is written as: the compensator gain is written as: the compensator transfer function is written as: figure 6. system control block diagram g vovc (s) g comp (s) v comp v o v ref t(s)=g comp (s) + g vovc (s) + ? figure 7. open loop transfer function 10 100 1 . 10 3 1 . 10 4 1 . 10 5 1 . 10 6 40 20 0 20 40 60 80 100 120 150 120 90 60 30 0 30 60 90 gain (gvovc) phase (gvovc) vcomp to vo transfer function gvovc(s) frequency (hz) gain (db) phase figure 8. closed loop transfer function 10 100 1 . 10 3 1 . 10 4 1 . 10 5 1 . 10 6 40 20 0 20 40 60 80 100 120 30 15 0 15 30 45 60 75 90 gain (tv) phase (tv) voltage loop gain t(s) frequency (hz) gain (db) phase z1 1 r comp c comp2 ? ------------------------------------------- - = (eq. 19) f z1 z1 2 ? --------- - = (eq. 20) p1 1 c comp1 --------------------- 1 c comp2 --------------------- + 1 r comp ------------------ ? = (eq. 21) f p1 p1 2 ? ---------- = (eq. 22) i 1 r comp c comp1 c comp2 + [] ? ---------------------------------------------------------------------------- - = (eq. 23) g comp s () i 1 s z1 --------- - + ? s1 s p1 ---------- + ? ---------------------------------- = (eq. 24) figure 9. system control block diagram + ? comp fb r comp c comp1 c comp2 r bottom v ref error amplifier ISL6269
17 all intersil u.s. products are manufactured, asse mbled and tested utilizing iso9000 quality systems. intersil corporation?s quality certifications ca n be viewed at www.intersil.com/design/quality intersil products are sold by description only. intersil corpor ation reserves the right to make changes in circuit design, soft ware and/or specifications at any time without notice. accordingly, the reader is cautioned to verify that data sheets are current before placing orders. information furnishe d by intersil is believed to be accurate and reliable. however, no responsibility is assumed by intersil or its subsidiaries for its use; nor for any infringements of paten ts or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of intersil or its subsidiari es. for information regarding intersil corporation and its products, see www.intersil.com fn9177.0 june 14, 2005 ISL6269 quad flat no-lead plastic package (qfn) micro lead frame pl astic package (mlfp) l16.4x4 16 lead quad flat no-lead plastic package (compliant to jedec mo-220-vggc issue c) symbol millimeters notes min nominal max a 0.80 0.90 1.00 - a1 - - 0.05 - a2 - - 1.00 9 a3 0.20 ref 9 b 0.23 0.28 0.35 5, 8 d 4.00 bsc - d1 3.75 bsc 9 d2 1.95 2.10 2.25 7, 8 e 4.00 bsc - e1 3.75 bsc 9 e2 1.95 2.10 2.25 7, 8 e 0.65 bsc - k0.25 - - - l 0.50 0.60 0.75 8 l1 - - 0.15 10 n162 nd 4 3 ne 4 3 p- -0.609 --129 rev. 5 5/04 notes: 1. dimensioning and tolerances conform to asme y14.5-1994. 2. n is the number of terminals. 3. nd and ne refer to the number of terminals on each d and e. 4. all dimensions are in millimeters. angles are in degrees. 5. dimension b applies to the meta llized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 6. the configuration of the pin #1 identifier is optional, but must be located within the zone indicated. the pin #1 identifier may be either a mold or mark feature. 7. dimensions d2 and e2 are fo r the exposed pads which provide improved electrical and thermal performance. 8. nominal dimensions are provided to assist with pcb land pattern design efforts, see intersil technical brief tb389. 9. features and dimensions a2, a3, d1, e1, p & are present when anvil singulation method is used and not present for saw singulation. 10. depending on the method of lead termination at the edge of the package, a maximum 0.15mm pull back (l1) maybe present. l minus l1 to be equal to or greater than 0.3mm.

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